GB1565899A - Creation of Inventions and Patents with Artificial Intelligence

GB1565899A – Circuit arrangemnt for receiving one of the sidebands of a double sideband signal
– Google Patents

GB1565899A – Circuit arrangemnt for receiving one of the sidebands of a double sideband signal
– Google Patents
Circuit arrangemnt for receiving one of the sidebands of a double sideband signal

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Publication number
GB1565899A

GB1565899A
GB5200077A
GB5200077A
GB1565899A
GB 1565899 A
GB1565899 A
GB 1565899A
GB 5200077 A
GB5200077 A
GB 5200077A
GB 5200077 A
GB5200077 A
GB 5200077A
GB 1565899 A
GB1565899 A
GB 1565899A
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United Kingdom
Prior art keywords
frequency
low
circuit
sideband
pass filter
Prior art date
1977-12-14
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)

Expired

Application number
GB5200077A
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Koninklijke Philips NV

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Philips Gloeilampenfabrieken NV
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
1977-12-14
Filing date
1977-12-14
Publication date
1980-04-23

1977-12-14
Application filed by Philips Gloeilampenfabrieken NV
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Philips Gloeilampenfabrieken NV

1977-12-14
Priority to GB5200077A
priority
Critical
patent/GB1565899A/en

1980-04-23
Publication of GB1565899A
publication
Critical
patent/GB1565899A/en

Status
Expired
legal-status
Critical
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Classifications

H—ELECTRICITY

H03—ELECTRONIC CIRCUITRY

H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER

H03D1/00—Demodulation of amplitude-modulated oscillations

H03D1/22—Homodyne or synchrodyne circuits

H03D1/2245—Homodyne or synchrodyne circuits using two quadrature channels

Description

(54) A CIRCUIT ENGAGEMENT FOR RECEIVING ONE
OF THE SIDEBANDS OF A DOUBLE SIDEBAND
SIGNAL
(71) We, N. V. PHILIPS’ GLOEIL
AMPENFABRIEKEN, a limited liability
Company, organised and established under the laws of the Kingdom of the Netherlands, of Emmasingel 29, Eindhoven, the
Netherlands, do hereby declare the invention, for which we pray that a patent may be granted to us, and the method by which it is to be performed, to be particularly described in and by the following statement:
The invention relates to an electric circuit arrangement for receiving one of the sidebands of a double sideband signal, which circuit arrangement is of a type comprising two first mixing stages in which the double sideband signal is mixed with a corresponding one of two locally produced radio frequency oscillations which are shifted in phase over 90″ relative to one another but are of the same frequency, the output signals of the two mixing stages being filtered by means of first and second lowpass filters, amplified and applied, respectively to two second mixing stages in which they are each mixed with a corresponding one of two locally produced low-frequency oscillations which are shifted in phase over 90″ relative to one another but are of the same frequency, the output signals of the two second mixing stages being applied to an adder (subtractor) circuit.
A circuit arrangement of the above type is known-especially in connection with a receiver for single sideband signals (cf., for example, DT-AS 16 16312 and “Proc. IRE 44” (1956) pages 1703 to 1705). The frequency of the locally produced radio frequency oscillation corresponds to the medium frequency of the received sideband and the frequency of the locally produced low-frequency oscillation corresponds to the medium frequency of the useful signal spectrum (third method). With this circuit there is the risk that the high frequency oscillation is radiated and – will cause squeaking in adjacent receivers. The second mixing stages in which the low-pass-filtered output signal of the preceding mixing stages is mixed with the low frequency oscillation and which are, for example, constituted as balanced modulators, must be of a very symmetrical construction as otherwise a distortion is produced in the centre of the useful frequency band owing to mixing with the low frequency oscillation. External screening and symmetry measures are therefore required, which partially offset again the advantages of this circuit arrangement (e.g. the fact they can be easily integrated, as coils are no longer required).
However, the invention relates to a circuit arrangement for receiving doublesideband signals, particularly with a full or an attenuated carrier, which can be utilized for example in medium-wave or short-wave broadcasting.
The prior art circuits generally untilize a frequency conversion for producing an intermediate frequency and require band filters in the intermediate frequency range to suppress signals from adjacent channels.
The known receiver circuits can therefore only be partially implemented in integrated circuit technology and the required filters must be connected externally. A further drawback of the known circuits consists in distortions when the so-called “selective fadings” occur. This effect, which is extremely annoying in radio reception, is produced by the fact that the carrier changes its amplitude or its phase position relative to the two sidebands. Consequently, a qualitatively satisfactory reception of the signals from remote medium-wave or shortwave transmitters is generally impossible.
Also the reception of the image frequency and the risk of direct reception on the intermediate frequency as well as the occurrence of spurious beats and squeaking in the receiving range are amongst the unpleasant properties of these circuits.
It is an object of the present invention to provide a circuit arrangement which enables improved reception of doublesideband AM-signals, also from remote transmitters, and which can be implemented in a simple manner in integrated circuit technology and which reduces spurious beats and squeaking.
According to the invention a circuit arrangement of the type referred to is characterised in that:
a) the frequency of the locally produced radio-frequency oscillation corresponds to the carrier frequency of the double-sideband signal,
b) the frequency of the locally produced low-frequency oscillation is greater than or equal to the upper frequency of the useful signal to be transmitted,
c) the adder (subtractor) circuit is followed by a third low-pass filter which suppresses signals having a frequency equal to or above the frequency of the locally produced low-frequency oscillation, and
d) the output signals of the third low-pass filter are mixed in a last mixing stage with the locally produced low-frequency oscillation.
An embodiment of the invention provides that the last mixing stage is followed by a fourth low-pass filter whose cutoff frequency corresponds to that of the third low-pass filter. By mixing the output signals of the third low-pass filter, with the low frequency oscillation in the last mixing stage, two sidebands are produced, only one of which (the lower one) can be used. It is true that the frequency of the low-frequency oscillation can be chosen so that the upper band of these two sidebands is outside the useful frequency range or outside the audio frequency range respectively, but the requirements imposed on the third low-pass filter and which separates the two sidebands of the double-sideband signal would be much higher.
Further, an embodiment of the invention ensures that, for the reception of doublesideband signals with carriers, the radio frequency oscillation is supplied by a tunable oscillatory circuit, whose frequency is synchronized by means of a phase and/or frequency control ciruit with the instantaneous carrier frequency. A so-called
PLL-circuit (Phase Locked Loop), which compares the frequency of the oscillator signal with the frequency of the carrier, comprised in the input signal, and which so changes the oscillator frequency that the difference of the phase position between receiver carrier and the locally produced oscillation is brought to a minimum, can be used as phase or frequency control circuit.
This also requires a low-pass filter having a very small or switch able cutoff frequency and a variable control voltage amplifier.
With the circuit arrangment according to the invention only one of the two sidebands comprised in the double-sideband signal is always used for reception. So a change in the phase position of the carrier relative to the two sidebands has no influence on the quality of the reception, which would manifest itself in a very disturbing manner as selective fading in the conventional double-side-band receivers. Separating the two sidebands from one another is effected in the third low-pass filter connected behind the adder (subtractor) unit. If very low frequencies, for example, 100 Hz or lower, must be transmitted in the two sidebands, the frequency gap between these two sidebands is then 200 Hz or less. This frequency gap also exists at the output of the adder (subtractor) circuit: however, as a result of the mixing of the double-sideband signal with the radio frequency and the low frequency oscillation it is moved to the low frequency range. The third low-pass filter, which separates the sidebands, must consequently satisfy very stringent requirements: It must have a passband corresponding to the bandwidth of the useful signal to be transmitted (some kHz) and the transition between the passband and the cutoff range should only be 100 Hz (or less). An embodiment of the invention therefore provides that this low-pass filter is constituted by a gyrator filter.
Also the other low-pass filters can be advantageously implemented as gyrator filters. For the first and second filters which follow the first radio frequency mixers, there is the additional condition that they must be fully identical as regards amplitude and also phase response. The phase of the oscillation presented to said filters by the two mixers is shifted in one branch 90″ relative to the other branch and this phase shift must be maintained in the filters over the entire frequency range.
Controlled amplifiers satisfy, as far as they are included in the two parallel branches, high requirements as regards equality of control behaviour. The required amplitude control voltage (AGC) is obtained by means of a low-pass filter of a very low cut-off frequency and a selfregulating control voltage amplifier. Any remaining amplitude errors in the two branches can then be compensated for by differential gain control in said two branches.
The invention will now be explained in greater detail with reference to an embodiment shown in the drawing by way of example. In the drawing: Figure 1 is a block diagram of a circuit arrangement according to the invention and
Figures 2a to 2g show the frequency spectra in various points of the circuit arrangement according to the invention.
The double-sideband signal u0 received from the antenna 1 is applied, -for example via broadband amplifiers, control amplifiers, attenuators, input circuits etc., which are not shown in the drawing-with equal phase to two identically constructed parallel branches A and B. Figure 2a shows the frequency spectrum of the signal uO. The signal contains a carrier of the carrier frequency fT and the two sidebands U and 0, which are arranged symmetrically at the carrier.
Each of the two branches comprises a first mixing stage la or lb a low-pass filter 2a or 2b and a further second mixing stage 3a or 3b. In the mixing stage la or lb the input signal u0 is mixed with a radio frequency oscillation supplied by a tunable oscillator 4, which oscillation is applied directly to the mixing stage la and via a phase shifting member 5, which shifts the locally produced oscillation over exactly 90″ to the mixing stage lb. Preferably the mixing stages la or lb are formed as fullwave multipliers supplying an output signal which is proportional to the product of the two input signals. In this way it is achieved that the output signal comprises only sideband informations whose frequencies deviate from the receiver carrier frequency.
It is important that the frequency of the oscillation supplied by oscillator 4 conforms as exactly as possible to the carrier frequency fT of the double-side-band signal uO. To this end a phase or frequency control circuit-indicated by the block 17, 18-is provided by means of which the frequency of the locally produced oscillation 4 is synchronized on the carrier frequency fT. To control the phase or frequency, the small positive or negative d.c. voltage which is produced in branch B in the case of phase errors, can be used. This voltage is applied via a low-pass filter 17 and the controlled amplifier 18 to the oscillator 4 and which readjust the tuning of the oscillator, so that the phase of the locally produced oscillation corresponds, except for a small residual error, to the portion of the carrier frequency comprised in the input signal uO.
Whilst the positive or negative d.c.
voltage occurring in branch B is proportional to the residual-phase angle + A 0, the d.c. voltage component in branch A is proportional to the instantaneous amplitude of the received carrier voltage and, after having been filtered by the low-pass filter 19 and gain-controlled in the controlled d.c.
voltage amplifier 20, can be used for amplitude control (AGC).
The d.c. voltage components are then removed by the capacitors 7a and 7b before the second mixers 3a and 3b.
It is evident that all known measures for linearizing, such as RF-feedbacks, highcurrent transistors, balanced circuits etc may be used for the mixers la and lb to avoid cross modulations. Furthermore by the mixing process of the input signals with the local high frequency oscillations the upper and lower sidebands of the received signal are so transposed to the lower frequency range that the two sidebands are superimposed, one sideband becoming, as it were, meshed on the other.
However, by mixing the signals with the two oscillator voltages whose phases are shifted over 90″ relative to one another, it is achieved that the signals in branch B have shifted over +90 for the upper band signal
O and over -90″ for the lower band signal
U, whereas in branch A the signals are not shifted and have a 0 phase shift. The signals u1 in branch A are therefore shifted over 90″ relative to those in branch B.
In the Figures 2b and 2c, which show the signals u, and U1′ respectively at the output of the low-pass filters 2a and 2b respectively, this situation is illustrated in that in Figure 2c the frequency spectrum originating from the lower sideband u is plotted with a negative sign, whereas in
Figure 2b the frequency spectrum originating from both sidebands is indicated with a positive sign.
If, after multiplication in the first mixer, the sideband oscillations 0 and U in branch
A are cosine functions, then the sideband oscillations produces in branch B are a sine and a minus-sine function.
This is important for the mixing processes in the second mixers, wherein a further 90″- shift is effected, so that 1800 is obtained.
The low-pass filter 2a and 2b respectively have for their object to remove the mixing product at or about twice the carrier frequency, produced at the output of the mixing stages la and lb. This filter furthermore serves for suppressing the mixing product produced by mixing the locally produced oscillation with the signal of a transmitter which is adjacent as regards frequency. If the frequency spacing of the carrier of two frequency-adjacent transmitters is, for example, 9 kHz-as prescribed in the plans of the CCIR- the cutoff frequency of the low-pass filter 2a or 2b must be half this spacing in the assumed example, consequently 4.5 kHz or less, so that the mixing products originating from adjacent transmitters are suppressed. The low-pass filters 2a or 2b are consequently used for channel or transmitter separation and they, consequently, have the same function as the intermediate frequency filter in a conventional superheterodyne receiver.
At the reception of double-sideband transmissions in which each sideband contains exactly the same information, the signal u, at the ouput of the low-pass filter 2a may already represent the low-frequency useful signal; however, the selective fading already indicated in the preamble, might be produced when the received carrier is not accurately symmetrical to the two sidebands or when the locally produced oscillation continuously changes its phase position relative to the carriers comprised in the input signal. The effects of the selective fading could only be removed by the subsequent portion of the circuit.
The output signal of each of the low-pass filters 2a and 2b respectively are supplied via a controlled amplifier 15 and 16 respectively to either of a further multiplicative mixing stage 3a and 3b respectively, in which it is mixed with the low-frequency oscillation produced by an oscillator 8, which oscillation is supplied directly to a mixing stage 3a and to the other mixing stage 3b through a phase shifting member 9, which shifts the phase of the locally produced oscillation over exactly 90″. The lowfrequency oscillation supplied by oscillator 8 must have a frequency fN, which is equal to or only slighty larger than, the upper frequency of the useful low-frequency signal to be transmitted; it must not be chosed too high, as the requirements on the low-pass filter separating the upper from the lower sideband and which will be explained in greater detail hereinbelow, are the greater according as the frequency of the locally produced oscillation is higher. If the frequency spacing between the carriers of two adjacent transmitters is, for example, 9 kHz, a value of 4.5 kHz, is then preferably choser in the frequency fN of oscillator 8.
The–preferably multiplicative-mixing stages fold the frequency spectra in accordance with Figure 2b and 2c repectively symmetrically on frequency fN of the added locally produced oscillation so that, for example, from the frequency band
O+U (Figure 2b), the frequency bands O+U and O’+ U’ are produced (cf. Figure 2d), and from the frequency bands 0 and U (Fig. 2c), the frequency bands 0, U and 0′, U’ (Fig.
2e). Those frequency components in the two sidebands which correspond to the low useful frequencies have a smaller frequency spacing from the frequency fN than the components which correspond to the higher useful frequencies. The second mixing operation with the two oscillations, which are shifted 90″ relative to one another, of the oscillator 8, furthermore accomplishes that the upper sudeband 0 appears in the lower frequency band (frequencies < fN) in the same phase position at the two outputs of the mixers 3a and 3b respectively, whereas the lower sideband U appears in the lower frequency band with the opposite phase at the two outputs of the mixer stages 3a and 3b (cf. Figure 2d and 2e). (The reverse situation is obtained if the phase shifting member 9 is not arranged between the oscillator 8 and the mixing stage 3b, but between the oscillator 8 and the mixing stage 3a). In the upper frequency band (i.e. for frequencies > fN) the lower sideband U’ is present, on the contrary, at the two outputs of the mixer stages in the same phase position, whereas the upper sideband 0′ is in the opposite phase position. This situation is also shown in
Figure 2d and Figure 2e.
The different phase positions of the sidebands U and 0 in the lower frequency band can be used for suppressing one of the sidebands. To this end an adder (subtractor) circuit 10 is provided to whose inputs the output signals u2 and u2′ of the two mixing stages 3a and 3b are supplied and which is preferably so constructed that the two signals may optionally either be added to or be subtracted from one another, so that either the lower or the upper sideband is suppressed in the lower frequency band. In the upper frequency band the other sideband is then suppressed. Figure 2f shows the output signal u3 of the circuit 10 after adding (subtracting). This output signal is applied to a low-pass filter I I, which suppresses all frequencie above the oscillator frequencyf,, as illustrated by the dashed representation of the upper frequency band. Consequently, only the original upper or the lower sideband- depending on whether the output signals of the mixing stages 3a and 3b are added or subtracted in the circuit 10, remains at the output of the low-pass filter. Unfortunately, the low frequency information is in the inverted frequency position so that a further mixing operation is required in mixer 13 for obtaining the correct position.
The above explanations will make it clear that the suppression of one sideband depends on as accurate as possible a compensation in the circuit 10 of the two components of the output signals of the mixer stages 3a and 3b corresponding to this sideband and on that the upper frequency band is suppressed as far as possible by the low-pass filter 11.
It is a requirement for the accurate compensation of either of the two sidebands that the two branches comprising the elements la, 2a, 3a, 15, 7a and respectively lb, 2b, 3b, 16, 7b are as identical as possible.
The best possible way to achieve this is by combining corresponding components in an integrated circuit.
To obtain a suppression of the upper frequency band which is as accurate as possible the low-pass filter 11 is preferably a gyrator-low-pass filter, which is constructed so, that its first pole position is substantially equal to the frequency of the oscillator 8. To this end a control circuit can be formed, comprising a phase comparison circuit 12 producing a signal which depends on the phase difference between the voltages at the input and the output of the low-pass filter 11, which signal is utilized for adjustment. Use is made of the fact that in the vicinity of the pole position of the attenuating filter this phase difference is highly dependent on the frequency. Instead of controlling the pole position of the low pass filter 11, it is also possible to control the frequency if the oscillator 8 by means of the phase comparison circuit 12. This is indicated in Fig. 1 by a dashed line.
The output signal of the low-pass filter 11 comprises the upper or the lower sideband in the inverted frequency position (inverted
LF-band), that is to say the higher frequency components of the sideband correspond to lower useful frequencies and the lower frequencies in the sideband correspond to higher useful frequencies. To bring the given sideband in the correct receiving position, a last mixing stage 13 is provided in which the output signal of the low-pass filter is mixed with the signal produced by oscillator 8. Last mentioned stage 13 must be-like the mixing stages 3a and 3b-a multiplicative mixing stage, for example a product detector, so that its output signal contains only the sum and difference frequencies formed from its two input signals (cf. Figure 2g). The sum frequencies then produced, which represent the given sideband U” or O” in the inverted position, are removed by means of a lowpass filter 14, whose cutoff frequency also corresponds to the frequency fN of oscillator 8. The output signal of last-mentioned lowpass filter represents the useful lowfrequency signal.
As mentioned above, the effects of the selective fading are avoided in the circuit arrangement according to the invention by using only one of the two sidebands U or O for reception. The adder (subtractor) circuit
10 can then be selected to operate in the mode (adding or subtracting) in which the less disturbed sideband is received. As only one sideband is used for reception, two sidebands which are independent from one another can also be received with this circuit arrangement.
As the frequency of oscillator 4 corresponds with exactly the same phase to the carrier frequency, it is impossible that, owing to the radiation of this frequency, interferences are produced, neither in the receiver comprising the circuit according to the invention, nor in the adjacent receivers.
A remainder of the frequency fN which may still be present in the output signal of the mixing stage 13 can be suppressed without further measures by the low-pass filter 14.
Coils are not required in the overall receiver so that the circuit can be implemented without further measures in integrated circuit technology. It is not required then to combine all the modules in one single integrated circuit, but it is important that modules which must have the same characteristic, for example the mixing stages 3a and 3b, are integrated on one and the same semiconductor substrate. In principle, it is alternatively possible to receive double sideband signals without carrier with the circuit according to the invention. In this case, however, a PLLcircuit cannot be used and the frequency of the oscillator 4 must very accurately correspond to the frequency of the suppressed carrier. In the most simple case the usual variably tuned input selectivity can be fully dispensed with, or it is possible to use input band filters which are fixedly adjusted to the receiving band.
WHAT WE CLAIM IS:- 1. A circuit arrangement for receiving one of the sidebands of a double sideband signal, comprising two first mixing stages in which the double sideband signal is mixed with a corresponding one of two locally produced radio frequency osciliations, which are shifted in place over 90″ relative to one another, but are of the same frequency, the output signals of the two mixing stages being filtered by means of first and second lowpass filters, amplified and applied, respectively, to two second mixing stages in which they are each mixed with a corresponding one of two locally produced low-frequency oscillations which are shifted in phase over 90″ relative to one another but are of the same frequency, the output signals of the two second mixing stages being applied to an adder (subtractor) circuit, characterized in that:
a) the frequency of the locally produced radio-frequency oscillation corresponds to the carrier frequency of the double sideband signal,
b) the frequency of the locally produced low-frequency oscillation is greater than or equal to the upper frequency of the useful signal to be transmitted,
c) the adder (subtractor) circuit is followed by a third low-pass filter which suppresses signals having a frequency equal to or above the frequency of the locally produced low frequency oscillation, and
d) the output signals of the third low-pass filter are mixed in a last mixing stage with the locally produced low frequency oscillation.
2. A circuit arrangement as claimed in
Claim 1, characterized in that the last
**WARNING** end of DESC field may overlap start of CLMS **.

Claims (13)

**WARNING** start of CLMS field may overlap end of DESC **.
constructed so, that its first pole position is substantially equal to the frequency of the oscillator 8. To this end a control circuit can be formed, comprising a phase comparison circuit 12 producing a signal which depends on the phase difference between the voltages at the input and the output of the low-pass filter 11, which signal is utilized for adjustment. Use is made of the fact that in the vicinity of the pole position of the attenuating filter this phase difference is highly dependent on the frequency. Instead of controlling the pole position of the low pass filter 11, it is also possible to control the frequency if the oscillator 8 by means of the phase comparison circuit 12. This is indicated in Fig. 1 by a dashed line.
The output signal of the low-pass filter 11 comprises the upper or the lower sideband in the inverted frequency position (inverted
LF-band), that is to say the higher frequency components of the sideband correspond to lower useful frequencies and the lower frequencies in the sideband correspond to higher useful frequencies. To bring the given sideband in the correct receiving position, a last mixing stage 13 is provided in which the output signal of the low-pass filter is mixed with the signal produced by oscillator 8. Last mentioned stage 13 must be-like the mixing stages 3a and 3b-a multiplicative mixing stage, for example a product detector, so that its output signal contains only the sum and difference frequencies formed from its two input signals (cf. Figure 2g). The sum frequencies then produced, which represent the given sideband U” or O” in the inverted position, are removed by means of a lowpass filter 14, whose cutoff frequency also corresponds to the frequency fN of oscillator 8. The output signal of last-mentioned lowpass filter represents the useful lowfrequency signal.
As mentioned above, the effects of the selective fading are avoided in the circuit arrangement according to the invention by using only one of the two sidebands U or O for reception. The adder (subtractor) circuit
10 can then be selected to operate in the mode (adding or subtracting) in which the less disturbed sideband is received. As only one sideband is used for reception, two sidebands which are independent from one another can also be received with this circuit arrangement.
As the frequency of oscillator 4 corresponds with exactly the same phase to the carrier frequency, it is impossible that, owing to the radiation of this frequency, interferences are produced, neither in the receiver comprising the circuit according to the invention, nor in the adjacent receivers.
A remainder of the frequency fN which may still be present in the output signal of the mixing stage 13 can be suppressed without further measures by the low-pass filter 14.
Coils are not required in the overall receiver so that the circuit can be implemented without further measures in integrated circuit technology. It is not required then to combine all the modules in one single integrated circuit, but it is important that modules which must have the same characteristic, for example the mixing stages 3a and 3b, are integrated on one and the same semiconductor substrate. In principle, it is alternatively possible to receive double sideband signals without carrier with the circuit according to the invention. In this case, however, a PLLcircuit cannot be used and the frequency of the oscillator 4 must very accurately correspond to the frequency of the suppressed carrier. In the most simple case the usual variably tuned input selectivity can be fully dispensed with, or it is possible to use input band filters which are fixedly adjusted to the receiving band.
WHAT WE CLAIM IS:- 1. A circuit arrangement for receiving one of the sidebands of a double sideband signal, comprising two first mixing stages in which the double sideband signal is mixed with a corresponding one of two locally produced radio frequency osciliations, which are shifted in place over 90″ relative to one another, but are of the same frequency, the output signals of the two mixing stages being filtered by means of first and second lowpass filters, amplified and applied, respectively, to two second mixing stages in which they are each mixed with a corresponding one of two locally produced low-frequency oscillations which are shifted in phase over 90″ relative to one another but are of the same frequency, the output signals of the two second mixing stages being applied to an adder (subtractor) circuit, characterized in that:
a) the frequency of the locally produced radio-frequency oscillation corresponds to the carrier frequency of the double sideband signal,
b) the frequency of the locally produced low-frequency oscillation is greater than or equal to the upper frequency of the useful signal to be transmitted,
c) the adder (subtractor) circuit is followed by a third low-pass filter which suppresses signals having a frequency equal to or above the frequency of the locally produced low frequency oscillation, and
d) the output signals of the third low-pass filter are mixed in a last mixing stage with the locally produced low frequency oscillation.

2. A circuit arrangement as claimed in
Claim 1, characterized in that the last
mixing stage is followed by a fourth low-pass filter whose cut-off frequency corresponds to that of the third low-pass filter.

3. A circuit arrangement as claimed in
Claim I or Claim 2 for receiving one from a plurality of double sideband signals whose carrier frequencies ae equally spaced (for example 9 kHz), especially for medium wave broadcast reception, characterized in that the frequency of the locally produced low-frequency oscillation corresponds to hald the frequency spacing of two adjacent carrier frequencies.

4. A circuit arrangement as claimed in any one of Claims 1 to 3 for receiving double-sideband signals with carriers, characterized in that the radio frequency oscillation is supplied by a tunable oscillator circuit, whose frequency is synchronized with the received carrier frequency by a phase and/or a frequency control circuit.

5. A circuit arrangement as claimed in
Claim 4, characterized in that the d.c.
voltage produced by the mixing stage whose input is supplied with the locally produced radio frequency oscillation with a phase shift of approximately 900 relative to the carrier oscillation, is used as control voltage for the phase control circuit.

6. A circuit arrangement as claimed in any one of Claims 1 to 5, characterized in that the d.c. voltage is produced by the mixing stage, whose input is supplied with the locally produced radio frequency oscillation having the same phase as the carrier oscillation, is used as control voltage for gain control.

7. A circuit arrangement as claimed in any one of Claims 1 to 4, characterized in that the third low-pass filter is constituted by a gyrator filter.

8. A circuit arrangement as claimed in any one of Claims I to 7, characterized in tha the first and second low-pass filters are constituted by gyrator filters.

9. A circuit arrangement as claimed in any one of Claims 2 to 8, characterized in that the fourth low-pass filter is a gyrator filter.

10. A circuit arrangement as claimed in any one of Claims 1 to 5, characterized in that a control circuit for adjusting the oscillator for locally producing the low frequency oscillation or for adjusting the first attenuation pole of the third low-pass filter is provided.

I 1. A circuit arrangement as claimed in any one of Claims 1 to 10, characterized in that the adder (subtractor) circuit is constructed so that the output signals of the second mixing stages can optionally be added or subtracted from one another.

12. A circuit arrangement as claimed in any one of Claims I to 11, characterized in that the d.c. components behind the first mixing stages are blocked by capacitors.

13. A circuit arrangement substantially as hereinbefore described with reference to the accompanying drawings.

GB5200077A
1977-12-14
1977-12-14
Circuit arrangemnt for receiving one of the sidebands of a double sideband signal

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Circuit arrangemnt for receiving one of the sidebands of a double sideband signal

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Circuit arrangemnt for receiving one of the sidebands of a double sideband signal

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Circuit arrangemnt for receiving one of the sidebands of a double sideband signal

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Cited By (4)

* Cited by examiner, † Cited by third party

Publication number
Priority date
Publication date
Assignee
Title

EP0184873A1
(en)

1984-11-30
1986-06-18
Koninklijke Philips Electronics N.V.
Phase-locked loop, particularly for use in a directly mixing synchronous AM receiver

EP0282607A1
(en)

1987-03-14
1988-09-21
Deutsche ITT Industries GmbH
Frequency-converter for television signal

AT394918B
(en)

1990-04-05
1992-07-27
Klaus Dipl Ing Dr Kerschbaumer

METHOD FOR DIRECTLY DEMODULATING AN RF SIGNAL

GB2328813A
(en)

1997-08-28
1999-03-03
Mitel Corp
RF zero IF direct converter system

1977

1977-12-14
GB
GB5200077A
patent/GB1565899A/en
not_active
Expired

Cited By (7)

* Cited by examiner, † Cited by third party

Publication number
Priority date
Publication date
Assignee
Title

EP0184873A1
(en)

1984-11-30
1986-06-18
Koninklijke Philips Electronics N.V.
Phase-locked loop, particularly for use in a directly mixing synchronous AM receiver

EP0282607A1
(en)

1987-03-14
1988-09-21
Deutsche ITT Industries GmbH
Frequency-converter for television signal

US4789897A
(en)

1987-03-14
1988-12-06
Deutsche Itt Industries Gmbh
Frequency converting apparatus for converting an RF television signal to a video signal employing low IF techniques

AT394918B
(en)

1990-04-05
1992-07-27
Klaus Dipl Ing Dr Kerschbaumer

METHOD FOR DIRECTLY DEMODULATING AN RF SIGNAL

GB2328813A
(en)

1997-08-28
1999-03-03
Mitel Corp
RF zero IF direct converter system

US6148184A
(en)

1997-08-28
2000-11-14
Mitel Corporation
Radio frequency zero if direct down converter

GB2328813B
(en)

1997-08-28
2001-08-29
Mitel Corp
A radio frequency zero IF direct down converter

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